Portable communication devices such as cellular-type telephones or other communication devices are becoming more widespread. A portable communication device includes one or more power amplifiers for amplifying the power of the signal to be transmitted from the portable communication device.
With the decreasing size of portable communication devices, power efficiency is one of the most important design criteria. Reducing power consumption prolongs power source life and extends stand-by and talk time of the portable communication device.
A portable communication device may employ a constant or a non-constant envelope modulation methodology. A non-constant envelope modulation scheme is typically implemented with a linear power amplifier. The entire amplitude and phase modulated waveform is provided to the input of the power amplifier and the power amplifier amplifies the combined signal. In a non-constant envelope modulation scheme, “power control” can be implemented as a “slow loop” regulating the gain of the power amplifier or adjusting the input amplitude to compensate for gain variation in the power amplifier that occurs due to process and temperature variations. Unfortunately, a linear power amplifier is significantly less efficient than a nonlinear power amplifier and, as such, consumes more power.
In the case where both a constant envelope modulation methodology and a non-constant envelope modulation methodology are employed, such as in a communication device that operates using the Global System for Mobile Communication (GSM) and the Enhanced Data Rates for GSM Evolution (EDGE) communication formats, the same power amplifier should be used for both signals. The GSM system provides a slightly higher output power and uses a constant-envelope modulation methodology. The EDGE system uses a non-constant-envelope modulation methodology. If a linear power amplifier is used to implement EDGE, then the power amplifier is less efficient when operated in GSM mode. This is why it is desirable to find a way to make a non-linear power amplifier work in EDGE mode.
Polar modulation is a known technique of performing non-constant envelope modulation using a nonlinear power amplifier. In polar modulation, a phase modulated input signal is applied to the radio frequency (RF) input to the power amplifier. The output power of the power amplifier is adjusted at the rate of the amplitude modulation to recompose the modulated waveform at the output of the power amplifier.
A GSM system has traditionally been implemented using a nonlinear power amplifier, with the “power control” implemented as a (slow) gain modulation in the power amplifier. A “power control” signal is supplied to the power amplifier from the baseband subsystem to implement the time-slotting (ramp up power at the beginning of the time slot, ramp it down at the end) of the communication protocol using this slow gain modulation. One prior attempt at implementing a power amplifier in the EDGE system using polar modulation increases the performance of the “power control” signal, so that the power amplifier output power can be changed rapidly to create the modulation and to create the power control (i.e. there is still the slow ramp up and ramp down at the edges of the slot, but the faster modulation is also added in the middle). In this manner, the power amplifier can still be used in GSM mode by applying a signal to the “power control” port with only the ramping signals, while also performing polar modulation in EDGE mode.
There are two kinds of polar modulation: open-loop and closed-loop. In open loop, there is no feedback path for the power amplifier output. In closed-loop, feedback on the amplitude and phase paths is used to measure the output amplitude and phase. The measured amplitude and phase are compared to a desired signal, and then an amplitude and gain correcting mechanism is used to minimize any discrepancy. Such an implementation is difficult while maintaining a very wide bandwidth, meeting noise requirements and preventing the system from becoming unstable and oscillating under output mismatch, for example, in the presence of a voltage standing wave ratio (VSWR).
In such a system, the phase modulation is typically applied directly to the signal input of the power amplifier. The phase can be controlled using a phase correction feedback loop.
One of the challenges when implementing a so called “polar modulation” technique is that the input to output phase relationship of the power amplifier can vary as the output power is changed to produce the amplitude modulation. This can induce phase error in the output signal, where the phase error is modulated with the amplitude modulation. This phase error can be reduced by using a phase correction feedback loop. One implementation of a phase correction feedback loop can use a phase shifter between the desired phase modulated signal and the signal input of the power amplifier, so that the phase shifter phase delay can be dynamically changed to compensate dynamically changing phase delays in the power amplifier.
In some other instances it is also desirable to alter the phase of an RF signal. Many high frequency RF applications use a controllable phase shifter. For example, such a phase shifter can be used in a phase modulator, phase shifting for the elements of a phased-array, and for phase correction loops such as when used with a phase detector and a feedback loop.
For many such applications, it is desirable that the amount of phase shift be continuously adjustable over a given range rater than in discrete steps. It is also often desirable that the amount of phase shift can be controlled over a wide range, possibly even more than 360 degrees. For instance, in a phase-correction feedback loop, large continuous phase shift range can allow the feedback loop to smoothly correct a large amount of power amplifier phase delay variation.
In many applications, it is also desirable that the amount of phase shift be controlled to be a linear function of the adjustment. For example, if the phase shift is adjusted using a control voltage, it is desirable that the amount of phase shift be approximately proportional to the control voltage (or proportional to the control voltage with an offset) so that the slope of the phase shift can be approximately constant across the control range. For instance, in a phase-correction feedback loop, the loop gain can be related to the slope of the phase shifter, so that changes in the phase shifter slope may adversely affect the behavior of the loop, such as by causing instability or reducing the effectiveness of the correction as the slope changes.
FIG. 1 is a schematic diagram of a prior art continuously-variable phase shifter 100 using a variable capacitor to adjust the phase shift. The phase shifter 100 uses a resistor 101 and a variable capacitor 102 connected so that the pole frequency of the transfer function from the input to the output can be modified by changing the capacitance value. The example in FIG. 1 depicts a resistor 101 connected in series with the signal path and a variable capacitor 102 connected in parallel with the output connection. However, other configurations can be used, such as having the capacitor 102 in series with the signal path and the resistor 101 in parallel with the output connection.
The variable capacitor 102 can be implemented using various methods. For example, a metal-oxide-semiconductor field effect transistor (MOSFET) capacitance, such as the gate capacitance of a MOSFET, can be used. A junction capacitor, such as a junction varactor can also be used. Other capacitances, such as micro-electro-mechanical system (MEMS) capacitors or mechanically tunable air-gap capacitors may also be used, so long as the capacitance value is continuously adjustable over a range.
The phase shifter 100 is widely used, but suffers from several limitations. The range of possible phase shift using such a circuit is limited to less than 90 degrees, which may be insufficient in many applications. Additionally, the amplitude of the output signal changes significantly with the phase shift. This amplitude change is unacceptable in many applications. For example, in the phase shifter 100, the input voltage and output voltage have nearly the same amplitude and phase when the capacitor 102 is set to a small value. However, as the capacitance is increased, the output voltage amplitude begins to drop as the phase delay increases. When the phase delay approaches 90 degrees, the output amplitude can be nearly zero.
The phase response of the phase shifter 100 can also be very nonlinear. For example, the phase shifter 100 typically has a phase shift of −atan(2*pi*R*C). Using a typical variable capacitor, such as a MOSFET capacitor or a junction varactor, the resulting phase shift response versus the control voltage can be very nonlinear.
FIG. 2 is a schematic diagram of a prior art continuously-variable phase shifter 200 using a variable resistor to adjust the phase shift. The phase shifter 200 uses a variable resistor 201 and a capacitor 202 connected so that the pole frequency of the transfer function from the input to the output can be modified by changing the resistance value. The example in FIG. 2 depicts a variable resistor 201 connected in series with the signal path and a capacitor 202 connected in parallel with the output. However, other configurations can be used, such as having the capacitor 202 connected in series with the signal path and the variable resistor 201 connected in parallel with the output.
The variable resistor 201 can be implemented using various methods. For example, a MOSFET can be used in the triode region such that the resistance value can be adjusted by changing the gate voltage of the MOSFET. Other resistor implementations may also be used, so long as the resistance value is continuously adjustable over a range.
The phase shifter 200 is widely used, but suffers from similar limitations as the variable capacitance phase shifter. The range of possible phase shift using such a circuit is limited to less than 90 degrees. Additionally, the amplitude of the output signal changes significantly with the phase shift, becoming nearly zero at one side of its phase shift range. The phase response versus resistance is similar to the phase shifter 100, typically being very nonlinear using typical variable resistances such as FETs.
FIG. 3 is a schematic diagram of a prior art continuously-variable multi-stage phase shifter 300. The phase shifter 300 uses variable resistors 301, 303 and 305 and capacitors 302, 304 and 306. The variable resistors 301, 303 and 305 and the capacitors 302, 304 and 306 are arranged so that adjusting the resistance values modifies the pole frequencies of the phase shifter 300, resulting in phase shift. Alternately, variable capacitances can be used instead of (or in addition to) variable resistors.
The phase shifter 300, by using several stages of components can potentially increase the range of possible phase shifts. For instance, the three stage circuit depicted can potentially produce a phase shift range of up to 270 degrees if the component values are selected properly. Unfortunately, the amplitude can vary over the range of phase shift, and the phase response can be very nonlinear. Additionally, the noise generated by the phase shifter 300 can be high due to the use of several resistors and signal attenuation.
FIG. 4 is a schematic diagram of a prior art continuously-variable multi-stage inductive-capacitive (LC) phase shifter 400. The phase shifter 400 uses inductors 401, 403 and 405 and capacitors 402, 404 and 406. The inductors 401, 403 and 405 and variable capacitors 402, 404 and 406 are arranged so that adjusting the capacitance values modifies the pole frequencies of the phase shifter 400, resulting in phase shift.
The phase shifter 400, by using several stages of components can also potentially increase the range of possible phase shift. For example, the three stage phase shifter 400 can potentially produce a phase shift range of up to 540 degrees if the component values are selected properly. The amplitude also can vary considerably over the control voltage, but due to the use of inductors instead of resistors, the phase shifter 400 can resemble a synthetic transmission line over a wide range of capacitance values and can have reasonably constant output amplitude over this range. Additionally, since the circuit uses no resistors, the noise induced by the phase shifter 400 can be quite low. Unfortunately, the phase shift can be very nonlinear over the control range, and the use of several inductors can be prohibitive for use in integrated circuit implementations, where resistors and capacitors typically consume less area.
Another prior art adjustable phase shifter generates several, typically two, fixed phase shifts using passive circuits including inductors and capacitors. These signals are added together in various proportions to produce phase shifted outputs as weighted vector averages of the fixed phase input signals. This prior art phase shifter also includes a circuit to set the gains of such a phase shifter having two paths, which can generate around 90 degrees of phase shift with reasonably constant amplitude output.
The prior art phase shifter uses independent LC ladder circuits to generate each phase shift. As such, it can be subject to errors and variation in each of the independent phase shifts due to component variation, resulting in unwanted slope variation in the phase shift response. Additionally, the use of several inductors can increase size and cost, particularly if the phase shifts are fabricated on an integrated circuit. Furthermore, although such a system using these phase shifters is capable of a system range of 180 degrees, it only discloses phase shifters capable of 90 degrees of range. The gain of each of the individual paths in this system can be difficult to control because the system exhibits gain as a function of transistor parameters as well as inductor values. This may cause unwanted variation in phase slope and output amplitude across the phase range if not carefully managed.
Therefore, it is desirable to have a phase shifter which can have larger than 90 degrees of control range with repeatable linear phase control gain and approximately constant amplitude which can be implemented in a compact integrated circuit.